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university:courses:electronics:text:chapter-14 [29 Sep 2013 20:48]
Doug Mercer created
university:courses:electronics:text:chapter-14 [06 Jun 2017 17:55]
Doug Mercer [14.3 Bandgap References]
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 +======Chapter 14:  Voltage References======
 +
 +=====14.1 Introduction =====
 +
 +Voltage references and linear voltage regulators have much in common. In fact, the latter could be 
 +functionally described as a reference circuit, but at a greater current (or power) output level. Accordingly, ​
 +many of the specifications of the two circuit types have great commonality (even though the output voltage ​
 +tolerance of references is usually tighter with regard to temperature drift, accuracy, etc.).
 +
 +Voltage references have a major impact on the performance and accuracy of analog systems. A ±5 mV tolerance ​
 +on a 5 V reference corresponds to ±0.1% absolute accuracy which is only 1 part in a thousand or 10-bit ​
 +accuracy. For a system with 12-bit accuracy, choosing a reference that has a ±1 mV tolerance may be far 
 +easier than manually calibrating a lower accuracy reference. Both high initial accuracy and calibration are 
 +likely to be necessary in a system making absolute 16-bit measurements. Note that many systems make relative ​
 +or ratiometric measurements rather than absolute ones. In such cases the absolute accuracy of the reference ​
 +is not as important, although noise and short-term stability may be.
 +
 +Temperature drift or drift due to aging may be an even greater problem than absolute accuracy. The initial ​
 +error can always be adjusted or calibrated, but compensating for drift over temperature or time is difficult. ​
 +Where possible, references should be designed for low temperature drift and aging characteristics which 
 +preserve adequate accuracy over the operating temperature range and expected lifetime of the system. Noise in 
 +voltage references is often overlooked, but it can be very important in system design. Noise is an 
 +instantaneous change in the reference voltage. It is generally specified on component data sheets, but system ​
 +designers frequently ignore the specification and mistakenly assume that their voltage reference does not 
 +contribute noise in their system.
 +
 +There are two dynamic issues that must be considered with voltage references: their behavior at start-up, and 
 +their behavior with transient loads. With regard to the first, always bear in mind that voltage references do 
 +not power up instantly (this is true of references inside power supply regulators, ADCs and DACs as well as 
 +discrete designs). Thus it is rarely possible to turn on an ADC and reference, whether internal or external, ​
 +make a reading, and turn off again within a relatively few microseconds,​ however attractive such a operation ​
 +might be in terms of energy saving.
 +
 +Regarding the second point, a given reference IC may or may not be well suited for pulse or fast transient ​
 +loading conditions, dependent upon the specific architecture. Many references use low power, and therefore ​
 +low bandwidth, output buffer amplifiers. This makes for poor behavior under fast transient loads, which may 
 +degrade the performance of high speed ADCs (especially capacitor based successive approximation and pipe-line ​
 +ADCs). Suitable decoupling can ease the problem (but some references oscillate with large capacitive loads), ​
 +or an additional external broadband buffer amplifier may be used to drive the part of the circuit where the 
 +transients occur.
 +
 +=====14.2 Simple Diode References =====
 +
 +In terms of the functionality of their circuit connection, standard reference ICs are often only available in 
 +series, or three-terminal form (V<​sub>​IN</​sub>,​ Common, V<​sub>​OUT</​sub>​),​ and also in positive polarity only. 
 +The series types have the potential advantages of lower and more stable quiescent current, standard ​
 +pre-trimmed output voltages, and relatively high output current without accuracy loss. Shunt, or two-terminal ​
 +(i.e. diode-like) references are more flexible regarding operating polarity, but they are also more 
 +restrictive as to loading. They can in fact eat up excessive power with widely varying resistor-fed voltage ​
 +inputs. Also, they sometimes come in non-standard voltages. All of these various factors tend to govern when 
 +one functional type is preferred over the other. By contrast, these most simple references (as well as all 
 +other shunt-type regulators) have a basic advantage, which is the fact that the polarity is readily ​
 +reversible by flipping connections and reversing the drive current. However, a basic limitation of all shunt 
 +regulators is that load current must always be less (usually significantly less) than the driving current.
 +
 +Some simple diode-based references are shown in Figure 14.1. In the first of these, a current driven forward ​
 +biased diode (or diode-connected transistor) produces a voltage, V<​sub>​f</​sub>​= V<​sub>​REF</​sub>​. While the 
 +junction drop is somewhat decoupled from the raw supply, it has numerous deficiencies as a reference. Among 
 +them are a strong temperature coefficient (TC) of about -0.3%/°C, some sensitivity to loading, and a rather ​
 +inflexible output voltage, it is generally only available in increments of 0.65V.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f1.png?​600 |}}
 +
 +<WRAP centeralign>​ (a) Forward-biased Diode (b) Zener (Avalanche) Diode </​WRAP>​
 +<WRAP centeralign>​ Figure 14.1: Simple two terminal Diode Reference Circuits </​WRAP>​
 +
 +The significant negative temperature coefficient of the forward biased emitter base junction of a diode 
 +connected transistor is dependent on V<​sub>​BE</​sub>​. To explore this let us first reexamine the 
 +V<​sub>​BE</​sub>​ equation.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e1.png?​150 |}}
 +
 +At first glance one sees that absolute temperature,​ T, appears in the equation and the quantity ​
 +V<​sub>​T</​sub>,​ the thermal voltage (kT/q), has a positive temperature coefficient. This would make 
 +V<​sub>​BE</​sub>​ have a positive temperature coefficient if I<​sub>​S,</​sub>​ the junction saturation current, was 
 +constant over temperature. I<​sub>​S</​sub>​in fact has a very strong temperature coefficient as seen in the 
 +following equation for I<​sub>​S</​sub>​.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e2.png?​220 |}}
 +
 +Note: E<​sub>​g</​sub>​is the energy gap of Silicon\\
 +M represents the temperature dependence of mobility
 +
 +If we use this equation for I<​sub>​S</​sub>​ and insert it in the V<​sub>​BE</​sub>​ equation and then differentiate ​
 +with respect to temperature we get the following relationship for a constant I<​sub>​C</​sub>​.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e3.png?​230 |}}
 +
 +Note:
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e4.png?​100 |}}
 + ​and ​
 +{{ :​university:​courses:​electronics:​text:​chptr14_e5.png?​100 |}}
 +
 +We can see this in the temperature simulation plot shown in figure 14.2 where the current through a diode 
 +connected NPN transistor is set to 1mA, 2mA, 5mA and 10mA.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f2.png?​650 |}}
 +
 +<WRAP centeralign>​ Figure 14.2 V<​sub>​BE</​sub>​ vs. Temperature at 1mA, 2mA, 5mA and 10mA </​WRAP>​
 +
 +The slope of the 10 mA line is slightly less negative than the 1 mA line. In fact all the V<​sub>​BE</​sub>​ vs. 
 +temperature lines would converge at absolute zero when projected back to the origin. The voltage they 
 +converge to is approximately the energy gap of Silicon or 1.2 volts. This property is not very helpful by 
 +itself but when combined with another property of BJTs can result in a low temperature coefficient reference ​
 +as we will discuss in section 14.3 on the Bandgap reference.
 +
 +The V<​sub>​BE</​sub>​ voltage of the simple diode connected transistor of figure 14.1(a) can be used to generate ​
 +a regulated current reference as well, as shown in figure 14.3. In this circuit the simple diode connection ​
 +around Q<​sub>​1</​sub>​is replaced by emitter follower Q<​sub>​2</​sub>​. The V<​sub>​BE</​sub>​ of Q<​sub>​1</​sub>​ is 
 +impressed across R<​sub>​2</​sub>​ and the resulting current flows through Q<​sub>​2</​sub>​ to become ​
 +I<​sub>​REF</​sub>,​ neglecting the base currents of Q<​sub>​1</​sub>​ and Q<​sub>​2</​sub>​.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f3.png?​600 |}}
 +
 +<WRAP centeralign>​ Figure 14.3 V<​sub>​BE</​sub>​ generated reference current. </​WRAP>​
 +
 +The resulting reference current, I<​sub>​REF</​sub>,​ will be equal to V<​sub>​BE</​sub>​ divided by R<​sub>​2</​sub> ​
 +and have the strong negative temperature coefficient as we saw in figure 14.2. This negative temperature ​
 +coefficient of the current is often referred to as CTAT or Complementary To Absolute Temperature. We can 
 +compensate for this negative temperature drift by summing this current with another current with an equally ​
 +strong positive temperature coefficient. Remembering back to Chapter 11 Section 10 where we discussed the 
 +Peaking Current source, we have a circuit block that has such a positive temperature drift. We referred to 
 +this as a PTAT or Proportional To Absolute Temperature current. In figure 14.4 we combine the circuit from 
 +figure 14.3 (Q<​sub>​1</​sub>,​ Q<​sub>​2</​sub>,​ R<​sub>​1</​sub>​and R<​sub>​2</​sub>​) on the right with the peaking ​
 +current source from figure 11.18 in Chapter 11 (Q<​sub>​3</​sub>,​ Q<​sub>​4</​sub>,​ R<​sub>​3</​sub>​and R<​sub>​4</​sub>​).
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f4.png?​600 |}}
 +
 +<WRAP centeralign>​ Figure 14.4 Combining CTAT and PTAT currents to make a constant I<​sub>​REF</​sub>​ </​WRAP>​
 +
 +If we set the PTAT and CTAT currents to be roughly equal (70uA) at some nominal temperature then the sum of 
 +the two currents (140uA) will be approximately flat vs. temperature as we see in the simulation plot in 
 +figure 14.5. 
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f5.png?​650 |}}
 +
 +<WRAP centeralign>​ Figure 14.5 Combined CTAT and PTAT current sources makes a constant </​WRAP>​
 +
 +It is also important to point out here that if the temperature independent I<​sub>​REF</​sub>​ current were 
 +applied to a low temperature coefficient resistor a more or less temperature independent voltage reference ​
 +would be the result. The need for a zero TCR resistor is not strictly required if all the resistors used in 
 +the circuit are at the same temperature and have identical TCR.
 +
 +==== 14.2.1 Zener diodes ====
 +
 +In the second circuit of figure 14.1(b), a zener or avalanche diode is used, and an appreciably higher output ​
 +voltage results. While true zener breakdown occurs below 5 V, avalanche breakdown occurs at higher voltages ​
 +and has a positive temperature coefficient. Note that diode reverse breakdown is referred to almost ​
 +universally today as zener, even though it is usually avalanche breakdown. With a D<​sub>​1</​sub>​ breakdown ​
 +voltage in the 5 to 8 V range, the net positive TC is approximately such that it equals the negative TC of 
 +forward-biased diode D<​sub>​2</​sub>,​ yielding a small net TC of 100 ppm/°C or less when operated at the proper ​
 +bias current. Combinations of such carefully chosen diodes formed the basis of the early single package ​
 +"​temperature-compensated zener" references, such as the 1N821-1N829 series. ​
 +
 +The temperature-compensated zener reference is limited in terms of initial accuracy, since the best TC 
 +combinations fall at odd voltages, such as the 1N829'​s 6.2 V. And, the scheme is also limited for loading, ​
 +since for best TC the diode current must be carefully controlled. Unlike a fundamentally lower voltage (<2 V) 
 +reference, zener diode based references must of necessity be driven from voltage sources appreciably higher ​
 +than 6 V levels, so this precludes operation of zener references from 5 V or lower system supplies. ​
 +
 +References based on low TC zener (avalanche) diodes also tend to be noisy, due to the basic noise of the 
 +breakdown mechanism. This has been improved greatly with monolithic zener types, as is described further in 
 +section 14.4.
 +
 +=====14.3 Bandgap References =====
 +
 +The development of voltage references with low output voltages (<5 V) based on the bandgap voltage of silicon ​
 +has led to the introduction of various integrated circuits with low temperature drift. The bandgap reference ​
 +technique is attractive in IC designs because of several reasons; among these are the relative simplicity, ​
 +and the avoidance of zeners and their noise. However, very important in these days of ever decreasing system ​
 +supplies is the fundamental fact that bandgap devices operate at low voltages down to 1.2V or less. Not only 
 +are they used for stand-alone IC references, but they are also used within the designs of many other linear ​
 +ICs such as ADCs and DACs. 
 +
 +To understand the underlying concept of the Bandgap reference we first need to explore an important ​
 +relationship involving bipolar transistors. Imagine that we have two identical transistors as shown in figure 14.6.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f6.png?​600 |}}
 +
 +<WRAP centeralign>​ Figure 14.6 A powerful relationship in circuit design </​WRAP>​
 +
 +Let's assume that each of these devices is provided with voltages at the base and that the collector voltage ​
 +(V+) is positive enough to avoid saturation, which is generally greater than V<​sub>​BE</​sub>​. In this 
 +experiment we will need to measure the two collector currents and the difference in base voltage. Since we 
 +can monitor the collector currents, it should be easy to constrain the range of the currents to a value where 
 +the collector current and base voltage are related by the now familiar relationship:​
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e6.png?​200 |}}
 +
 +Where, I<​sub>​C</​sub>​ is the collector current, V<​sub>​BE</​sub>​ is the base-emitter voltage, I<​sub>​S</​sub>​ is 
 +the saturation current for the transistor with a particular geometry and doping, and q/kT is the reciprocal ​
 +of the thermal voltage. A fairly standard transistor operating at around 100uA may have a V<​sub>​BE</​sub>​of ​
 +around 650mV at room temperature where q/kT is about 0.039/V. The exponential factor in the equation will be 
 +on the order of 10<​sup>​11</​sup>​. In this case we can safely drop the -1 term without serious error. Using 
 +this approximation we can now investigate the effect of operating the matched transistors at different ​
 +currents. If we establish the two collector currents I<​sub>​C1</​sub>​ and I<​sub>​C2</​sub>​ by adjusting ​
 +V<​sub>​BE1</​sub>​ and V<​sub>​BE2</​sub>,​ then we can take the ratio of the two currents and remembering that 
 +I<​sub>​S1</​sub>​ is equal to I<​sub>​S2</​sub>,​ as follows by rearranging the equation:
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e7.png?​200 |}}
 +
 +From this equation we get the expected result that V<​sub>​BE1</​sub>​-V<​sub>​BE2</​sub>​=0 when the ratio of 
 +I<​sub>​C1</​sub>​ to I<​sub>​C2</​sub>​ is equal to 1. We can get the equation for the difference in V<​sub>​BE</​sub> ​
 +(ΔV<​sub>​BE</​sub>​) by taking the natural logarithm:
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e8.png?​350 |}}
 +
 +From these equations we can see that the I<​sub>​S</​sub>​ terms are gone so the strong negative temperature ​
 +effect is now gone as well. So the ΔV<​sub>​BE</​sub>​ now has a positive temperature coefficient and is in fact 
 +proportional to absolute temperature (PTAT).
 +
 +We can now proceed to the next step in developing a voltage reference. We now extend this concept for 
 +transistors with equal base voltages but different emitter areas as shown in figure 14.7. The V<​sub>​BE</​sub> ​
 +of Q<​sub>​1</​sub>​equals the V<​sub>​BE</​sub>​ of Q<​sub>​2</​sub>,​ which results in a controlled current ratio 
 +between I<​sub>​C1</​sub>​ and I<​sub>​C2</​sub>​ of 1:8 based on their relative emitter areas.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f7.png?​600 |}}
 +
 +<WRAP centeralign>​ Figure 14.7 Different emitter areas results in controlled current ratio </​WRAP>​
 +
 +We can now reduce the current in Q<​sub>​2</​sub>​back to be equal to that of Q<​sub>​1</​sub>​ by inserting a 
 +resistor between the emitter of Q<​sub>​2</​sub>​ and ground as in figure 14.8. I<​sub>​C2</​sub>​ x R<​sub>​1</​sub> ​
 +reduces the V<​sub>​BE</​sub>​ of Q<​sub>​2</​sub>​ changing the I<​sub>​C1</​sub>​ / I<​sub>​C2</​sub>​ ratio. The voltage ​
 +drop across the resistor, R<​sub>​1</​sub>​ represents the ΔV<​sub>​BE</​sub>​ at that particular current level. For 
 +a given value for R<​sub>​1</​sub>,​ there will be one and only one value of V<​sub>​BE</​sub>​where the two 
 +collector currents are equal (other than I<​sub>​C1</​sub>​=I<​sub>​C2</​sub>​=0). This is shown in the simulation ​
 +plot in figure 14.9 where for an emitter area ratio of 8 and a resistor value of 200 ohms.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f8.png?​600 |}}
 +
 +<WRAP centeralign>​ Figure 14.8 Inserting emitter resistor R<​sub>​1</​sub>​ </​WRAP>​
 +
 +At low currents where the voltage drop across R<​sub>​1</​sub>​ is relatively small, I<​sub>​C2</​sub>​ increases ​
 +more or less exponentially at 8 times I<​sub>​C1</​sub>​. As the voltage drop across R<​sub>​1</​sub>​ increases, the 
 +current I<​sub>​C2</​sub>​ becomes less and less exponential and eventually the still exponential I<​sub>​C1</​sub> ​
 +catches up and passes I<​sub>​C2</​sub>​. ​
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f9.png?​650 |}}
 +
 +<WRAP centeralign>​ Figure 14.9 Plot of collector current vs. V<​sub>​BE</​sub>​ </​WRAP>​
 +
 +If we can configure a circuit, usually through negative feedback, that adjusts V<​sub>​BE</​sub>​ in figure 14.8 
 +such that I<​sub>​C1</​sub>​=I<​sub>​C2</​sub>​ we can obtain a controlled value for ?​V<​sub>​BE</​sub>​. ​
 +
 +The basic principle of the Bandgap reference is examined here using the circuit originally proposed by Robert ​
 +Widlar in 1971 and is shown in figure 14.10. The fundamental idea Widlar used was to compensate the negative ​
 +TC of the base emitter voltage V<​sub>​BE</​sub>​ by summing it with a second voltage V(R<​sub>​2</​sub>​) which has 
 +a positive TC.
 +
 +All Bandgap references use two basic elements:\\
 +1. Two BJT's working at different current densities\\
 +2. Adding a V<​sub>​BE</​sub>​ (-TC) and a PTAT voltage drop (+TC)
 +
 +The problem is that in order to compensate the large negative TC of V<​sub>​BE</​sub>​ a rather large 
 +ΔV<​sub>​BE</​sub>​ on the order of 600mV would be required. This cannot be done with the simple circuit of 
 +figure 14.8.
 +
 +The first of these reference circuits was the LM109, and a basic bandgap cell is shown in Figure 14.10.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f10.png?​600 |}}
 +
 +<WRAP centeralign>​ Figure 14.10: Basic Bandgap Reference </​WRAP>​
 +
 +This circuit is also called a "​ΔV<​sub>​BE</​sub>"​ reference. The differing current densities between matched ​
 +transistors Q<​sub>​1</​sub>​-Q<​sub>​2</​sub>​ produce a ΔV<​sub>​BE</​sub>​ across R<​sub>​3</​sub>​. Because resistors ​
 +R<​sub>​1</​sub>​ and R<​sub>​2</​sub>​ are in the ratio of 1:10 the current in Q<​sub>​2</​sub>​ will be 1/10 that in 
 +Q<​sub>​1</​sub>​ resulting in a smaller V<​sub>​BE</​sub>​ for Q<​sub>​2</​sub>​. The output, V<​sub>​Ref</​sub>,​ is 
 +produced by summing the V<​sub>​BE</​sub>​ of Q<​sub>​3</​sub>​ with the amplified ΔV<​sub>​BE</​sub>​ of 
 +Q<​sub>​1</​sub>​-Q<​sub>​2</​sub>,​ developed across R<​sub>​2</​sub>​. The ΔV<​sub>​BE</​sub>​ and V<​sub>​BE</​sub> ​
 +components have opposite polarity TCs; ΔV<​sub>​BE</​sub>​ is proportional-to-absolute-temperature (PTAT), while 
 +V<​sub>​BE</​sub>​ is complementary-to-absolute-temperature (CTAT). When the summed output, V<​sub>​Ref,</​sub>​ is 
 +equal to 1.205 V (silicon bandgap voltage), the TC is a minimum. I<​sub>​IN</​sub>​ must be larger than the sum 
 +of I<​sub>​C1</​sub>​ and I<​sub>​C2</​sub>​ and the excess current will flow in Q<​sub>​3</​sub>​ as I<​sub>​C3</​sub>​.
 +
 +However, the basic designs of figure 14.10 suffer from load and current drive sensitivity,​ plus the fact that 
 +the output needs accurate scaling to more useful levels, i.e., 2.5 V, 5 V, etc. The load drive issue is best 
 +addressed with the use of a buffer amplifier, which can also provide convenient voltage scaling to standard ​
 +levels.
 +
 +An improved three-terminal bandgap reference, (the AD580 introduced in 1974) is shown in figure 14.11. ​
 +Popularly called the "​Brokaw Cell", this circuit provides on-chip output buffering, which allows good drive 
 +capability and standard output voltage scaling.
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f11.png?​600 |}}
 +
 +<WRAP centeralign>​ Figure 14.11: Brokaw Cell Precision Bandgap Reference (AD580 1974) </​WRAP>​
 +
 +The Brokaw Cell based AD580 was the first precision bandgap based IC reference, and variants of the topology ​
 +have influenced further generations of industry standard references.
 +
 +The popular design choice is two 8:1 emitter-scaled transistors Q<​sub>​1</​sub>​-Q<​sub>​2</​sub>​ operating at 
 +identical collector currents (and thus 1/8 current densities), by virtue of matched load resistors ​
 +R<​sub>​3</​sub>,​R<​sub>​4</​sub>​ and a closed loop feedback around the buffer op amp. Due to the resultant smaller ​
 +V<​sub>​BE</​sub>​ of the 8× area Q<​sub>​1</​sub>,​ R<​sub>​2</​sub>​ in series with Q<​sub>​1</​sub>​ drops the 
 +ΔV<​sub>​BE</​sub>​ voltage, while R<​sub>​1</​sub>​ (due to the current relationships) has a multiplied PTAT voltage ​
 +V<​sub>​R1</​sub>:​
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_e9.png?​150 |}}
 +{{ :​university:​courses:​electronics:​text:​chptr14_e10.png?​150 |}}
 +
 +The bandgap cell reference voltage V<​sub>​BG</​sub>​ appears at the combined base of Q<​sub>​1</​sub>​ and 
 +Q<​sub>​2</​sub>,​ and is the sum of V<​sub>​BE</​sub>​ (Q<​sub>​2</​sub>​) and V<​sub>​R1</​sub>,​ or 1.205 V, the bandgap ​
 +voltage:
 +
 +However, because of the presence of the R<​sub>​5</​sub>/​R<​sub>​6</​sub>​ resistor divider and the op amp, the 
 +actual voltage appearing at V<​sub>​OUT</​sub>​ can be scaled higher, in this case 2.5 V. Following this general ​
 +principle, V<​sub>​OUT</​sub>​ can be raised to almost any other practical level, such as for example with 
 +selectable taps for precise 2.5, 5, 7.5, and 10 V output values. The buffer amplifier can often provide up to 
 +10 mA output current while operating from supplies between 4.5 and 30 V. These kinds of references can have 
 +output tolerances as low as 0.4%, with TCs as low as 10 ppm/°C.
 +
 +**ADALM1000 Lab Activity 9, [[university:​courses:​alm1k:​alm-lab-9|Bandgap reference]]**\\
 +**ADALM1000 Lab Activity 10, [[university:​courses:​alm1k:​alm-lab-10|Shunt Bandgap reference]]**
 +
 +**ADALM2000 Lab Activity 9, [[university:​courses:​electronics:​electronics-lab-9|Bandgap reference]]**\\
 +**ADALM2000 Lab Activity 10, [[university:​courses:​electronics:​electronics-lab-10|Shunt Bandgap reference]]**
 +
 +=====14.4 Buried (sub-surface) Zener References=====
 +
 +In terms of the design approaches used within the reference core, the two most popular basic types of IC 
 +references consist of the bandgap and buried zener approaches. Bandgaps have been discussed, but zener based 
 +references warrant some further discussion.  ​
 + 
 +In an IC chip, surface operated diode junction breakdown is prone to crystal imperfections and other 
 +contamination,​ thus zener diodes formed at the surface are more noisy and less stable than are buried (or 
 +sub-surface) ones (see figure 14.17). Analog Devices'​ zener-based IC references employ the much preferred ​
 +buried zener. This improves substantially upon the noise and drift of surface-mode operated zeners (see 
 +Reference 4).  ​
 + 
 +Buried zener references offer very low temperature drift, down to the 1-2 ppm/°C (AD588 and AD586), and the 
 +lowest noise as a percent of full-scale, i.e., 100 nV/√Hz or less. On the downside, the operating ​
 +current of zener type references is usually relatively high, typically on the order of several mA. The zener 
 +voltage is also relatively high, typically on the order of 5V. This limit it's application in low voltage ​
 +circuits. A block diagram of the AD586 is shown in Figure 15.8. 
 + 
 +{{ :​university:​courses:​electronics:​text:​chptr14_f12.png?​500 |}}
 +
 +<WRAP centeralign>​ Figure 14.17: Simple Surface Zener vs. a Buried Zener </​WRAP>​
 +
 +{{ :​university:​courses:​electronics:​text:​chptr14_f13.png?​500 |}}
 +
 +<WRAP centeralign>​ Figure 14.18: Typical Buried Zener Reference (AD586) </​WRAP>​
 + 
 +An important general point arises when comparing noise performance of different references. The best way to 
 +do this is to compare the ratio of the noise (within a given bandwidth) to the dc output voltage. For 
 +example, a 10 V reference with a 100 nV/ sqrt Hz noise density is 6 dB more quiet in relative terms than is a 
 +5 V reference with the same noise level. ​
 +
 +**Return to [[university:​courses:​electronics:​text:​chapter-13|Previous Chapter]]**
 +
 +**Go to [[university:​courses:​electronics:​text:​chapter-15|Next Chapter]]**
 +
 +**Return to [[university:​courses:​electronics:​text:​electronics-toc|Table of Contents]]**
  
university/courses/electronics/text/chapter-14.txt · Last modified: 06 Jun 2017 17:55 by Doug Mercer